As known, integrated circuits comprising on a single silicon chip a plurality of operational amplifiers, which sometimes are internally connected to RC networks so as to implement integrator circuits, are commercially available. These circuits can be advantageously used in the design of integrated filters, since the number of external components required is very limited. More particularly, capacitors and external inductors can be avoided by the use of suitable techniques, e.g. of "Active R" type.
However, there are a number of disadvantages in the industrial use of such types of filters. A first disadvantage is due to the limited accuracy of the individual integrated elements, namely, transistors, resistors, capacitors, owing to unavoidable fabrication tolerances. As a consequence, the gain-bandwidth (G*B) product of the amplifiers or of the integrators of an integrated circuit is generally different from that of another integrated circuit. Since the filter cut-off frequency depends on that product, filters different from the computed ones can be obtained. Hence it was necessary to provide at least a tuning element outside the integrated circuit, upon which a skilled technician could act during an initial adjusting phase. Yet this kind of operation is costly and requires a more complicated integrated circuit, since suitable pins, connected to internal test points should be provided, although these pins will be no longer used after tuning. This problem will become even more serious if the filter belongs to a rather complex system, e.g. a modem.
Another disadvantage is due to gain-bandwidth product variations dependent on temperature variations, affecting bias current ratios, integrated capacities, transistor parameters, et cetera. Of course this disadvantage cannot be avoided by only occasional adjustments; but rather an automatic control system is required.
The desired filter should require neither initial tunings, whatever the integrated circuit employed, nor adjustments during the operation, owing to thermal drift. Its cut-off frequency ought to be certainly determined a priori and should not depend on the integrated-circuit characteristics.
Some systems for automatically controlling the gain-bandwidth product are already known in the literature. More particularly two systems of this kind are described in the article entitled "Continuous-Time MOSFET-C Filters in VLSI" by Yannis Tsividis et alii, IEEE Journal of solid-state circuits, Vol. SC-21, No. 1, February 1986, pages 15-29 and shown in FIGS. 1(b) and 1(c).
These are indirect methods, i.e. methods in which the control of gain-bandwidth product is carried out on one or more amplifiers on the same silicon chip, carrying the amplifiers actually used to implement the filter. Undergoing the same phases of the technological process, fabrication tolerances are the same, and being in close proximity on the chip, temperature variations are common to all the amplifiers. One or more amplifiers can then be used to measure G*B and to extract a signal proportional to it, which controls G*B of all the amplifiers present in the same integrated circuit. More particularly this error signal can be used to control bias currents upon which the amplifier G*Bs can depend.
These methods are also called "indirect tuning" methods, since, by controlling G*Bs, filter cut-off frequencies are controlled; these filters hence become tunable at the desired frequencies on the basis of a previous programmation.
According to the method of the system shown in FIG. 1(b) of the cited article, at least two operational amplifiers are used to implement a reference filter of the "biquad" type, to whose input a predetermined-frequency clock signal is sent. Said signal is also sent to a comparison circuit, which compares its phase with the phase of the same signal extracted at the filter output. An error signal is obtained from the comparison, which acts on the filter amplifiers to keep the phase difference at the chosen frequency at a predetermined and constant value, thereby compensating for fabrication tolerances and temperature variations. This method requires at the input a sinusoidal signal, which generally presents some difficulties in an environment in which digital signals are found and occupies at least two amplifiers and other elements for implementing the reference filter. Besides the method demands a four-quadrant analog multiplier for implementing the comparison circuit. As is known, the design of this multiplier presents considerable difficulties due to circuit complexity, since non-linearity introduced by transistors should be avoided.
The relation between the G*B product and the signal frequency at the reference-filter input is also rather difficult to compute.
According to the method of the system shown in FIG. 1(c), at least two amplifiers are interconnected to obtain a voltage-controlled oscillator. The signal produced is compared in a phase-comparator with a reference signal coming from the outside and the error signal, duly filtered, is used to stabilize the integrated-oscillator frequency. Thus a well-known "phase-locked loop", or PLL, is formed. Since the fabrication tolerances of the integrated circuit which comprises the amplifiers and temperature variations determine corresponding variations in the frequency of the signal generated by the voltage-controlled oscillator, its correction based on the frequency of the external signal causes the correction of the G*B products of all the amplifiers. This system unfortunately requires a further filter to limit residual ripple of error signal, and hence a further external capacitor. In fact it is not advisable to obtain this operation from the loop-filter alone, implementing it with very-low cut-off frequency, as locking difficulties could arise. The whole system ought to be designed as well as possible so as to always ensure integrated oscillator locking to the reference phase, otherwise G*B product control might be lost. This system also requires a circuit for the control of the signal level at the output of the voltage-controlled oscillator, in order to prevent it from being blocked at one of the power-supply voltage levels (+Vcc, -Vcc) owing to input voltage drift.